Battery monitor

ABSTRACT

A circuit tracks the total amount of time that a host device has spent in its high power “activated” state (an optionally idle and hibernate states) and thereby can estimate the total power consumed by the tag. A remote device can query the state of a counter storing a value representing this time to accurately determine how much of the energy has been consumed by the host device and how much time is left and/or how many more operations can be performed before the host device&#39;s battery is exhausted.

RELATED APPLICATIONS

The present invention is related to U.S. patent application Ser. No.______, entitled “ACCURATE PERSISTENT NODES” and filed concurrentlyherewith, and which is incorporated by reference.

FIELD OF THE INVENTION

The present invention relates to power consumption monitoring circuitry,and more particularly, this invention relates to circuits that monitorpower consumption in a host device.

BACKGROUND OF THE INVENTION

Radio Frequency Identification (RFID) technology employs a radiofrequency (“RF”) wireless link and ultra-small embedded computercircuitry. RFID technology allows physical objects to be identified andtracked via these wireless “tags”. It functions like a bar code thatcommunicates to the reader automatically without requiring manualline-of-sight scanning or singulation of the objects. RFID promises toradically transform the retail, pharmaceutical, military, andtransportation industries.

Several advantages of RFID technology are summarized in Table 1:

TABLE 1 Identification without visual contact Able to read/write Able tostore information in tag Information can be renewed anytime Unique itemidentification Can withstand harsh environment Reusable HighFlexibility/Value

As shown in FIG. 1, a basic RFID system 100 includes a tag 102, a reader104, and an optional server 106. The tag 102 includes an integratedcircuit (IC) chip and an antenna. The IC chip includes a digital decoderneeded to execute the computer commands the tag 102 receives from thetag reader 104. The IC chip also includes a power supply circuit toextract and regulate power from the RF reader; a detector to decodesignals from the reader; a back-scattering modulator to send data backto the reader; anti-collision protocol circuits; and at least enoughEEPROM memory to store its EPC code.

Communication begins with a reader 104 sending out signals to find thetag 102. When the radio wave hits the tag 102 and the tag 102 recognizesthe reader's signal, the reader 104 decodes the data programmed into thetag 102. The information can then be passed to a server 106 forprocessing, storage, and/or propagation to another computing device. Bytagging a variety of items, information about the nature and location ofgoods can be known instantly and automatically.

The system uses reflected or “backscattered” radio frequency (RF) wavesto transmit information from the tag 102 to the reader 104. Sincepassive (Class-1 and Class-2) tags get all of their power from thereader signal, the tags are only powered when in the beam of the reader104.

The Auto ID Center EPC-Compliant tag classes are set forth below:

Class-1

-   -   Identity tags (RF user programmable, maximum range ˜3 m)

Class-2

-   -   Memory tags (8 bits to 128 Mbits programmable at maximum ˜3 m        range)    -   Security & privacy protection

Class-3

-   -   Battery tags (256 bits to 64 Kb)    -   Self-Powered Backscatter (internal clock, sensor interface        support)    -   ˜100 meter range

Class-4

-   -   Active tags    -   Active transmission (permits tag-speaks-first operating modes)    -   Up to 30,000 meter range

In RFID systems where passive receivers (i.e., Class-1 tags) are able tocapture enough energy from the transmitted RF to power the device, nobatteries are necessary. In systems where distance prevents powering adevice in this manner, an alternative power source must be used. Forthese “alternate” systems (also known as active or semi-passive),batteries are the most common form of power. This greatly increases readrange, and the reliability of tag reads, because the tag doesn't needpower from the reader. Class-3 tags only need a 10 mV signal from thereader in comparison to the 500 mV that a Class-1 tag needs to operate.This 2,500:1 reduction in power requirement permits Class-3 tags tooperate out to a distance of 100 meters or more compared with a Class-1range of only about 3 meters.

One concern with powered tags is the life of the battery.Battery-powered RFID tags draw very little power when silent, but draworders of magnitude more power when active. If the tag has been forcedinto activated states many times, the battery will be used up morequickly than a tag activated less. Because certain tags are active moreoften than others, it is hard to estimate the battery life of a tag. Thecurrent method is to replace all tags when the battery on one of thetags dies, as it is likely others will die soon as well. However, manyother tags may still have a long life remaining. Thus it would bedesirable to estimate the remaining life of a battery in an RFID tag.

SUMMARY OF THE INVENTION

A circuit tracks the total amount of time that a host device such as anRFID tag has spent in its high power “activated” state (an optionallyidle and hibernate states) and thereby can estimate the total powerconsumed by the tag. A remote device such as an RFID reader can querythe state of a counter storing a value representing this time toaccurately determine how much of the energy has been consumed by the tagand how much time is left and/or how many more operations can beperformed before the tag's battery is exhausted. The circuit includesoscillators wherein the speed (e.g., frequency) of the oscillators iscontrolled by reference currents and current mirrors. The circuits mayalso include fixed-frequency ultra-low power oscillators with multipleor variable frequency dividers that can divide the fixed-frequencyoutput by varying amounts depending on whether or not the tag is in ahigher-power activate state, a lower-power inactive state, or otherpower state. The battery monitor circuit is designed to consume only asmall fraction of the power of the chip itself so as not tosignificantly shorten the battery life of the tag.

In one embodiment, a counter circuit includes an ultra low poweroscillator that initiates when the tag wakes up, runs when the tag isactive, and stops when the tag turns off. The oscillator count is thenadded to a value in a register. The reader can interrogate the registerstoring the count and estimate how much life is left in the battery,such as by comparing the count to a benchmark count.

Optionally, the oscillator can run during the tag's inactive states,thereby keeping track of total time.

In another embodiment, a second oscillator is run during tag inactivestate to estimate off-time usage. Note that this may require a secondregister to store the counts from the second oscillator. Then the readercan query the registers and estimate the remaining battery life usingboth active and inactive times.

In a further embodiment, the system has one clock that is run at twodifferent speeds, or a varying speed based on power consumption. In theformer case, the clock runs at a fast speed when the tag is active andat a slow speed when the tag is inactive. In the latter case, the clockruns proportional to power dissipation. The more power being used, thefaster the clock cycle (higher count). When the tag is idling orhibernating, the clock runs more slowly. In any case, the reader cancompare the total count to a benchmark value representing tag life. Forinstance, the reader can use an experimental average life count of, say7 million, and compare it to the total count in the tag to estimateremaining battery life.

Other aspects and advantages of the present invention will becomeapparent from the following detailed description, which, when taken inconjunction with the drawings, illustrate by way of example theprinciples of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

For a fuller understanding of the nature and advantages of the presentinvention, as well as the preferred mode of use, reference should bemade to the following detailed description read in conjunction with theaccompanying drawings.

FIG. 1 is a system diagram of an RFID system.

FIG. 2 is a system diagram for an integrated circuit (IC) chip forimplementation in an RFID tag.

FIG. 3 is a system diagram of a system that provides intermittentbattery monitoring.

FIG. 4 is a circuit diagram of a circuit for continuous batterymonitoring.

FIG. 5 is a circuit diagram of a circuit for continuous batterymonitoring with automatic calibration.

BEST MODE FOR CARRYING OUT THE INVENTION

The following description is the best embodiment presently contemplatedfor carrying out the present invention. This description is made for thepurpose of illustrating the general principles of the present inventionand is not meant to limit the inventive concepts claimed herein.

The following specification describes systems and methods for trackingthe total amount of time that a host device spends in a high-power“Active” state and optionally, low-power state(s), to thereby allowestimation of the total power consumed by the host device. Active statescan include those states where the device is actively using power, orwhere power consumption is above what would be considered a low-powerstate. Note that the host device can have multiple active states, andoperation of the system can vary for each active state. Low power statestypically include “hibernate”, “idle”, or “wait/listen for an activationcommand” states. A remote device can query the device to accuratelydetermine how much of the battery energy has been consumed by the hostdevice, how much life is left in the battery, and/or how many moreoperations can be performed before the host device's battery isexhausted.

Many types of devices can take advantage of the embodiments disclosedherein, including but not limited to Radio Frequency Identification(RFID) systems and other wireless devices/systems; pacemakers; portableelectronic devices; remote controllers for televisions, audio devices,and other electronic devices; smoke detectors; etc. To provide acontext, and to aid in understanding the embodiments of the invention,much of the present description shall be presented in terms of an RFIDsystem such as that shown in FIG. 1. It should be kept in mind that thisis done by way of example only, and the invention is not to be limitedto RFID systems, as one skilled in the art will appreciate how toimplement the teachings herein into electronics devices in hardwareand/or software. Examples of hardware include Application SpecificIntegrated Circuits (ASICs), printed circuits, monolithic circuits,reconfigurable hardware such as Field Programmable Gate Arrays (FPGAs),etc. Further, the methodology disclosed herein can also be incorporatedinto a computer program product, such as a computer disc containingsoftware. Further, such software can be downloadable or otherwisetransferable from one computing device to another via network,nonvolatile memory device, etc.

A preferred embodiment is preferably implemented in a Class-3 or higherClass RFID tag, but will function with any type of module or class ofRFID tag. FIG. 2 depicts a circuit layout of a Class-3 module 200according to a preferred embodiment for implementation in an RFID tag,and is presented by way of example only. This Class-3 module can formthe core of RFID modules appropriate for many applications such asidentification of pallets, cartons, containers, vehicles, or anythingwhere a range of more than 3 meters is desired. As shown, the module 200includes several industry-standard circuits including a power generationand regulation circuit 202, a digital command decoder and controlcircuit 204, a sensor interface module 206, a C1V2 interface protocolcircuit 208, and a power source (battery) 210. A display driver module212 can be added to drive a display.

A battery activation circuit 214 is also present to act as a wake-uptrigger. The battery activation circuit 214 includes with anultra-low-power, narrow-bandwidth preamplifier. The battery activationcircuit 214 also includes a self-clocking interrupt circuit and may usean innovative 32-bit user-programmable digital wake-up code as describedin U.S. patent application entitled “BATTERY ACTIVATION CIRCUIT” andhaving Ser. No. 11/007,973, filed on Dec. 8, 2004, and which is hereinincorporated by reference. The battery activation circuit 214 draws lesspower during its sleeping state and is much better protected againstboth accidental and malicious false wake-up trigger events thatotherwise would lead to pre-mature exhaustion of the Class-3 tag battery210.

A battery monitor 215 monitors power usage in the device. Theinformation collected can then be used to estimate a useful remaininglife of the battery.

A forward link AM decoder 216 uses a simplified phase-lock-looposcillator that requires an absolute minimum amount of chip area.Preferably, the circuit 216 requires only a minimum string of referencepulses.

A backscatter modulator block 218 preferably increases the backscattermodulation depth to more than 50%.

A pure, Fowler-Nordheim direct-tunneling-through-oxide mechanism 220 ispresent to reduce both the WRITE and ERASE currents to less than 0.1μA/cell in the EEPROM memory array. This will permit designing of tagsto operate at maximum range even when WRITE and ERASE operations arebeing performed.

The module 200 also incorporates a highly-simplified, yet veryeffective, security encryption circuit 222 as described in U.S. patentapplication entitled “SECURITY SYSTEM AND METHOD” and having Ser. No.10/902,683, filed on Jul. 28, 2004 and which is herein incorporated byreference.

Sensors to monitor temperature, shock, tampering, etc. can be added byappending an industry-standard I2C interface to the core chip.

Extremely low-cost Class-2 security devices can be built by simplydisabling or removing the wake-up module, pre-amplifiers, and IF modulesfrom the Class-3 module core.

Several embodiments are described below with reference to specificspeeds (frequencies), voltages, amperages, etc. to exemplify thepreferred low power aspects of the embodiments. It should be kept inmind that these are provided by way of example only, and any suitablespeed (frequency), voltage, amperage, etc. can be used, as will beapparent to one skilled in the art.

Intermittent Battery Monitoring:

FIG. 3 illustrates one embodiment of a system 300 having an RFID tagbattery monitor 302 in which a mechanism, e.g., dedicated oscillator304, tracks the total amount of time that the tag has spent in itshigh-power “Activated” state. Another mechanism, e.g., counter 306,generates a value based on the tracking, where the value can thereby beused to estimate the total power consumed by the tag. A remote devicesuch as a reader (see FIG. 1) can query the state of this counter 306 toaccurately determine how much of the battery energy has been consumed bythe tag, how much life is left in the battery and/or how many moreoperations can be performed before the tag's battery is exhausted.

In a simple embodiment, the battery monitor oscillator 304 operates onlywhen the tag is active. In this circuit 300, the battery monitoroscillator 304 is, using a frequency divider 312, divided down by theratio of 16:1 from the tag's internal oscillator 308 (typically a 40 KHzoscillator that might be used in the C1G2 specification) that the taguses to parse and decode data it receives from the reader. In this way,the battery monitor oscillator 304 operates accurately at 2.5 KHz sincethis frequency is ultimately derived from the reader's oscillator 310 oraccurately generated on the RFID chip to an accuracy of better than±15%. If the tag is commanded by the reader to download data at a fasterrate than 40 Kbps, then the tag may increase the divider ratio for thebattery monitor 302 above 16:1 (as shown in FIG. 3) to make sure thebattery monitor oscillator 304 continues to run at 2.5 KHz. In this way,the battery monitor oscillator 304 can provide a consistent tracking ofthe battery usage, regardless of data speed.

The battery monitor oscillator 304 drives the “Battery Monitor Counter”306 that increments when the oscillator 304 is running and continues tostore the current cumulative count during periods when the tag isinactive or unpowered. The contents of the battery monitor counter 306can be read in a pre-assigned location of the optional user memory bank314 of the tag. In this example, the oscillator 304 drives the counter306 at 2.5 KHz. The counter 306 is preferably at least 32-bits long tomake sure it never overruns its storage limit, but only the state of the16 MSBs have to be addressable by the reader.

The divider 312 and the resulting slow 2.5 KHz oscillation frequency ofthe battery monitor oscillator 304 ensure that the power dissipation ofthe battery monitor circuit 302 is negligible compared to the rest ofthe active power dissipation of the tag. Typical power dissipation willbe only a few nanoamperes.

Those skilled in the art may replace the simple digital frequencydivider 312 with current mirrors and reference current sources. In asimilar manner, the battery monitor 302 need not operate at a fixedfrequency but may be made to vary in proportion with the variations inthe power dissipation of the RFID chip assuming that the powerdissipation may vary as a function of the forward data rate, whether thetag is writing data to memory or not, whether or not the tag isoperating a sensor, etc.

In a variation on the above, a second oscillator can be run during othertag states to estimate off-time usage. Note that this may require asecond register to store the counts from the second oscillator. Then thereader can query both registers and estimate the remaining battery lifeusing both active and inactive times.

Continuous Battery Monitoring with Analog Control:

In a more complex illustrative embodiment shown in FIG. 4, a tag batterymonitor 400 may also include a fixed-frequency ultra-low-poweroscillator 402 with multiple or variable frequency dividers that candivide the fixed-frequency output by varying amounts depending onwhether or not the tag is in a higher-power activate state, a low-powerinactive state, or other power state. As in the circuit 302 of FIG. 3,the battery monitor circuit 400 of FIG. 4 is designed to consume only asmall fraction of the power of the chip itself so as not tosignificantly shorten the battery life of the tag.

In this illustrative embodiment, assume the RFID tag contains atemperature sensor and supports 3 different operating modes:

-   -   a hibernate or “inactive” mode in which it consumes only 0.1 μA        in power;    -   an “activated” mode in which it exchanges data with the reader        and consumes 10 μA    -   a “sense” mode in which it takes and records a temperature        reading and consumes 30 μA

While the amount of time the tag will spend in each of these modesvaries widely, the circuit 400 monitors the time spent in each mode andaccumulates the effect this will have on battery life.

The circuit shown in FIG. 4 contains a precision calibratedUltra-Low-Power (“ULP”) oscillator 402 that runs at a frequency rate ofonly about 500 Hz and consumes only about 3 nA of power. This oscillator402 can run continuously which permits more accurate measurement ofpower consumption since it now monitors and measures inactive“hibernate” power consumption in addition to the active powerconsumption measured in FIG. 3. Addition of the ULP oscillator 402 alsofacilitates other important tag functions like enabling a real timeclock, enabling logging of temperature and other sensor data, etc.

The ULP oscillator 402 in this embodiment consists of a VoltageControlled Oscillator (“VCO”) formed with three inverters (I1, I2, I3)connected with feedback to form a ring oscillator. The frequency of thisoscillator is controlled by matched pairs of current mirroredtransistors P5/N5, P6/N6, P7/N7, and a capacitor connected to the outputof I1. The current flowing in these transistors is in turn controlledboth by the “2 nA reference current” flowing in P3 and the analogcurrent multiplier circuit formed by N1, N2, N3, and N4.

In the lowest-power “Inactive” mode, the tag is neither in the “Sense”nor “Active” state, and the negative-sense inputs “NSense” and “NActive”are both high. This effectively connects transistors N1, N2, and N3 inparallel at the drain of N3 for a total conductivity of 1×+2×+297×=300×.The “1×” or “297×” here refers to the relative “size”, “conductivity”,or more accurately “g_(m)” of the transistors. Following standardcurrent-mirror design practice, the effective mirror ratio is now30×/300× which sets the currents flowing through P4 and N4 (plus eachthe other current mirrored transistor pairs P5/N5, P6/N6, and P7/N7) at0.2 nA. An additional 2 pf of low-leakage non-junction capacitance isalso added to the output of inverter I1 to reduce the oscillatorfrequency to about 500 Hz. Note that in this “inactive” mode the powerdissipation of this entire battery monitoring circuit totals only about5 nA, which is much less than the power consumed by the tag itself.

However, when the tag enters the “active” state, transistor N2 isdisconnected from the mirror circuit. This changes the currentmultiplier ratios as follows: the total conductivity is now 3×(1×+2×),the ratio is now 30×/3×, the P4 current has increased to 20 nA, and theVCO oscillator frequency has increased from 500 Hz to 50 KHz. Note thatwhen the tag power dissipation increased 100× from 0.1 AA to 10 μA, thebattery monitor circuit responded by increasing the counter frequency by100× also. Also, note that although the monitor power dissipationincreased to about 0.1 μA, it remains less than 1% of the total powerdissipation of the chip itself.

Finally, when the tag enters the highest-power “sensor” state,transistors N2 and N3 are both disconnected from the mirror circuit.This changes the current multiplier ratios as follows: the totalconductivity is now only 1×, the ratio is now 30×/1×, the P4 current hasincreased to 60 nA, and the VCO oscillator frequency has increased to150 KHz. Note that when the tag power dissipation increased 300× from0.1 μA to 30 μA, the battery monitor circuit responded by increasing thecounter frequency by 300× also. Also, note that although the monitorpower dissipation increased to about 0.3 μA, it remains less than 1% ofthe total power dissipation of the chip itself. It is well known tothose skilled in the state of the art that well-designed VCO oscillatorslike the one shown in FIG. 4 or described in U.S. Pat. No. 4,236,199,can be accurately controlled over frequency ranges of 10,000:1 or more.

A Calibrated Ultra-Low-Power Current Source:

While the preceding discussion shows how the battery monitor circuitwill work with a 2 nA reference current source, no such current sourceshave ever existed in the IC chip world. For example, just trying toscale a conventional PMOS transistor to source only this much current(with it's gate grounded and it's source at 1.2V) would require thechannel length to be scaled to over 100,000 microns—hardly a practicaldesign. And in any case, the accuracy and stability of any 2 nA currentsource would be extremely poor without a method for accuratelycalibrating this current. FIG. 4 therefore includes a practical circuitfor generating and calibrating the 2 nA current-source. Again, it shouldbe stressed that the 2 nA current source specification is by way ofexample only, and higher and lower currents can be used with thecircuit.

In FIG. 4, the calibrated current source is controlled by a replica ofthe first VCO that is used to drive the accumulating counter 404, exceptthat this reference VCO 406 runs at a constant frequency independentlyof the Activate or Sense modes of operation. In this example the VCO 406operates at 5000 Hz, but again, it could operate at either a higher orlower frequency. The output of the reference VCO 406 is buffered andclipped to form a square wave and used to drive P1 and P2. P1, P2 andtheir associated capacitors form a “switched capacitor” precisionresistor 408. The bias current flowing through this network isnominally: I=(C1)×(f)×(ΔV)=20 ff×500 Hz×0.5V=50 pA. Nominally, this 50pA bias current also flows through the calibration matrix and induces anoffset voltage across the calibration matrix of 0.75V—assuming that 3 ofthe bypass calibration transistors are turned off. With a nominal Pthreshold voltage of 0.44V, then both P3 and P13 will be biased atexactly 0.01 V above their thresholds and will in theory each injectexactly 2 nA into the circuit.

In practice P3 and P13 (and the other mirrored transistors in thecircuit) can operate either just above or below their respectivethreshold voltages, i.e. both can be operated in their sub-thresholdregion if necessary to keep the reference current low. Also in practicenone of these nominal variables are well controlled, so withoutcalibration, the resulting reference currents and oscillationfrequencies might vary greatly from their nominal values due tovariations in threshold voltage, temperature, and sub-thresholdcharacteristics of both the diodes and transistors.

Accurate calibration of the current source and the oscillator frequencyis therefore achieved as follows. The reader issues a “Calibration”command and sends a 5000 Hz reference tone to the tag. The tag uses asimple PLL circuit 412 to compare the reader reference frequency withit's own reference oscillator frequency and adjusts the 5 digital inputsto the calibration matrix as necessary to force it's own referenceoscillator frequency to match that of the reader. Once set, the digitalcalibration settings are permanently stored in memory, e.g., eitherEEPROM or static RAM until a reader tells the tag to re-calibrateitself.

The calibration matrix shown in FIG. 4 is digitally adjustable. Thenominal maximum voltage across the calibration matrices is 1.25 V (250mV forward bias at 50 pA for each “1×-sized” diode). Coarse adjustmentsof about 250 mV are made by shorting out completely one or more of the“Cal” diodes. As shown in FIG. 5, the circuit of FIG. 4 can be extendedas necessary to provide digital adjustments as fine as 1 mV.

Continuous Battery Monitoring with Digital Control:

FIG. 5 shows another embodiment 500 of the invention including a singlecalibrated ULP oscillator 502 running at 1 KHz and two digital dividers504, 506 to monitor time in each of the three illustrative tag powermodes (Inactive, Active, Sensor) and to total the cumulative impact ofoperating in each of these modes on the remaining battery capacity. Whenoperating in the “Inactive” mode the 1 KHz oscillator output is divided300:1 before the output is fed into the 40-bit accumulating counter 508.In the “Active” mode the reference oscillator is divided 3:1, and in the“Sensor” mode it passes directly to the accumulating counter 508. Inthis circuit 500, changes to the reference oscillator frequency areminimized, and a constant 10 pA bias current passes through P1 and P2 tothe calibration matrix.

Auto-Calibration:

The calibration matrix shown in FIG. 5 is digitally adjustable with acombination of both fine and coarse adjustment bits. In this example,use of an ultra-low-bias current of only about 10 pA reduces the offsetvoltage across the parallel combinations of the four diodes (a,b,c,dwhich total 22.1× in size) to only about 200 mV. In one preferredembodiment, the tag will calibrate itself with the following simplifiedalgorithm.

Initially all 24 of the calibration transistors are turned on and thecalibration matrix is shorted out completely. The tag referencefrequency will then initially exceed that of the reader referencefrequency and this fact is detected by the PLL oscillator 510 shown inFIG. 5. In response, the calibration logic starts turning off each ofthe N1 a, N2 a, N3 a . . . transistors in sequence until the PLL detectsthat the tag frequency has dropped below that of the reader referenceoscillator (or until N1 a through N6 a are all off). Each disconnected“a-series” transistor increases the voltage to the gate of P3 by 200 mV.If and when the PLL detects that the tag frequency has dropped below thereader reference frequency of 1000 Hz, the calibration circuit turns thelast two “a-series” transistors that it had switched off, back on. The“a-series” coarse calibration sequence is now complete.

Next, the tag begins the “b-series” calibration sequence by turning offeach of the N1 b, N2 b . . . transistors in sequence until the PLLdetects that the tag frequency has dropped below that of the readerreference oscillator (or until N1 b through N6 b are all off). Eachdisconnected “b-series” transistor decreases the size of the diode from22.18× to 2.18×, and this increases the voltage to the gate of P3 by 60mV. This is because the forward-current/junction-area of the diode is anexponential function of the forward voltage with a slope of about 60mV/decade at room temperature. If and when the PLL detects that the tagfrequency has dropped below the reader reference frequency of 1000 Hz,the calibration circuit turns the last two “b-series” transistors thatit had switched off, back on. The “b-series” calibration sequence is nowcomplete.

Next, the tag begins the “c-series” calibration sequence by turning offeach of the N1 c, N2 c . . . transistors in sequence until the PLLdetects that the tag frequency has dropped below that of the readerreference oscillator (or until N1 c through N6 c are all off). Eachdisconnected “c-series” transistor decreases the size of the diode from2.18× to 1.18×, and this increases the voltage to the gate of P3 by 20mV, based on the equation ΔV=(log₁₀2.1/1.18)(60 mv/decade)=20 mV. Asbefore, if and when the PLL detects that the tag frequency has droppedbelow the reader reference frequency of 1000 Hz, the calibration circuitturns the last “c-series” transistor that it had switched off, back on.The “c-series” calibration sequence is now complete.

Finally, the tag begins the “d-series” calibration sequence by turningoff each of the N1 d, N2 d . . . transistors in sequence until the PLLdetects that the tag frequency has dropped below that of the readerreference oscillator. Each disconnected “d-series” transistor decreasesthe size of the diode from 1.18× to 1.0×, and this increases the voltageto the gate of P3 by 5 mV, based on the following equation:

ΔV=(log₁₀1.1/1.0)(60 mv/decade)=5 mV

The calibration circuit then stops and locks the digital inputs to eachof the calibration transistors in either EEPROM or static memory untilthe tag receives another “Calibration” command from the reader. The fullauto-calibration sequence is now complete.

If necessary, even finer adjustments could be made by connecting evenmore diodes of different sizes in parallel thereby controlling theforward drop by increments as small as 1 mV. The net effect is to adjustthe voltage across the calibration matrix such that at the nominal 10 pAbias current, there is just the right combination of diodes so that theinput voltage to P3 is exactly what is necessary to produce the 2 nAreference current. The negative feedback employed during the calibrationsequence ensures the tag will calibrate itself accurately despite thevariability in threshold voltages, leakage currents, etc.

Once calibrated, the bias voltage on the gate of P3 is maintained bynegative feedback through P1 and P2. If, for example, the P3 gatevoltage were to decrease, then the current through P3 would increase andthe reference frequency would also increase. This would increase thecurrent flowing through P1 and P2 which would fully restore the P3 gatevoltage to the original value set by the calibration sequence.

While circuits like those shown in FIGS. 4 and 5 can achieve initialfrequency and current calibration accuracies of better than ±10%, thisaccuracy may be degraded by changes in temperature or by subsequentvariations in the power supply voltage. The best results are achieved byminimizing the variability of the power supply voltage using either aband-gap regulator or a battery power supply. In addition, the regulatedpower supply could compensate for both the temperature effect on thethreshold voltage of P3/P13 and for the 2 mV/degree variation of thecalibration diodes. If necessary, the accuracy can also be furtherimproved by periodic re-calibration of the tag.

Additional Considerations:

To estimate the remaining life of the tag, the reader can query the tagfor the value stored in the counter (or derivative of the value) andcompare that value (or derivative thereof) to a benchmark valuerepresenting tag life or battery life. For instance, the reader can usean experimental average life count of, say 7 million, and compare it tothe value in the counter to estimate power usage and remaining batterylife.

If the user changes the tag battery, the count can be reset from thereader. Or the count can automatically reset if a battery is removed andreplaced. The reader can send an alert by email, integrated displayscreen, computer link, etc. if a tag life is nearing its end. Similarly,the tag can include an integrated display screen, activated at userrequest, to show an estimated power consumption/life remaining.

Note that the method and circuit herein could also apply to measuringthe service life of the tag, measuring tag activity, estimating aremaining useful life of the tag (e.g., if the circuitry is prone to alimited lifetime), etc.

There have thus been described circuits and systems that permit accuratemonitoring of the total charge consumed by an IC chip and thereby permitone to accurately predict the end of life of the battery. In addition,there has been described an accurate calibrated ULP oscillator that canoperate continuously at power levels as low as 3 nA, and frequencies inthe range of 10 Hz to 100 KHz without the use of external capacitors orother external components.

While various embodiments have been described above, it should beunderstood that they have been presented by way of example only, and notlimitation. Thus, the breadth and scope of a preferred embodiment shouldnot be limited by any of the above-described exemplary embodiments, butshould be defined only in accordance with the following claims and theirequivalents.

1. A system for estimating power consumption of a host device,comprising: a mechanism for tracking at least an amount of time a hostdevice is in an active state; a mechanism for generating a valuerepresenting at least the amount of time the host device is in theactive state based on the tracking; and a mechanism for sending thevalue or derivative thereof to a remote device.
 2. The system of claim1, wherein the mechanism for tracking an amount of time the host deviceis in the active state is a first oscillator.
 3. The system of claim 2,wherein the first oscillator operates at a fraction of a rate of aprimary oscillator of the host device.
 4. The system of claim 3, whereinthe first oscillator operates at no greater than one sixteenth the rateof the primary oscillator of the host device.
 5. The system of claim 2,wherein the first oscillator is idle when the host device is in aninactive state.
 6. The system of claim 2, wherein the first oscillatoris a voltage controlled oscillator, a frequency of the first oscillatorvarying depending on a power dissipation of the host device.
 7. Thesystem of claim 2, wherein a frequency of the first oscillator iscalibrated.
 8. The system of claim 1, wherein a rate that the valueincreases varies independently of a data transmission rate of the hostdevice, wherein the mechanism for tracking the amount of time isdependent upon a rate of a primary oscillator of the host device.
 9. Thesystem of claim 1, wherein a rate that the value increases variesdepending on a power dissipation rate of the host device.
 10. The systemof claim 1, wherein the mechanism for generating the value representingthe amount of time the host device is in the active state is a counter,the value being a cumulative count stored in the counter.
 11. The systemof claim 10, wherein the speed of the counter is digitally controlled.12. The system of claim 1, wherein the mechanism for tracking an amountof time the host device is in the active state is an oscillator, theoscillator operating at about a constant frequency, wherein at least onefrequency divider operates between the oscillator and the mechanism forgenerating the value.
 13. The system of claim 12, wherein the oscillatoroperates on a current of less than about one microampere (μA).
 14. Thesystem of claim 12, wherein the oscillator operates on a current of lessthan about 5-nanoamperes (nA).
 15. The system of claim 1, wherein theremote device uses the value to estimate usage of power in the hostdevice.
 16. The system of claim 1, wherein the system is embodied on aRadio Frequency Identification (RFID) tag.
 17. A method for estimatingpower consumption, comprising: tracking an amount of time a host deviceis in an active state; generating a value representing the amount oftime the host device is in the active state; and sending the value to aremote device.
 18. The method of claim 17, wherein the remote deviceuses the value to estimate power consumption in the host device.
 19. Themethod of claim 17, wherein the method is performed by a radio frequencyidentification (RFID) system.
 20. A system for estimating powerconsumption of a host device, comprising: an oscillator for tracking atleast an amount of time a host device is in a first state; a counterreceiving output from the oscillator for generating a value representinga power consumption of the host device based on output from theoscillator, wherein the counter operates at different speeds dependingon the state of the host device.
 21. The system of claim 20, wherein thehost device has a battery, wherein the host device calculates anestimation of a remaining battery life of the host device based on thepower consumption value.
 22. The system of claim 20, wherein the hostdevice has a battery, wherein a remote device calculates an estimationof a remaining battery life of the host device based on the powerconsumption value.
 23. The system of claim 20, wherein the frequency ofthe oscillator is digitally controlled.
 24. The system of claim 20,wherein the oscillator operates on a current of less than about onemicroampere (μA).
 25. The system of claim 20, wherein the oscillatoroperates on a current of less than about 5 nanoamperes (nA).
 26. Thesystem of claim 20, wherein a frequency of the oscillator is aboutconstant, wherein at least one frequency divider operates between theoscillator and the mechanism for generating the value.
 27. The system ofclaim 20, wherein a frequency of the oscillator varies depending on thestate of the host device.
 28. The system of claim 20, wherein the systemis embodied on a Radio Frequency Identification (RFID) tag.
 29. A methodfor estimating power consumption of a host device, comprising:generating a first tracking signal at a first frequency when a hostdevice is in a first state; generating a second tracking signal at asecond frequency when the host device is in a second state differentthan the first state; and processing the tracking signals for generatinga value representing a power consumption of the host device.
 30. Amethod for estimating power consumption of a host device, comprising:generating a tracking signal at a constant frequency when a host deviceis in a first or second state; dividing the frequency of the trackingsignal when the host device is in the second state; and storing a countof the frequency.
 31. A method for calibrating a current by linking thecurrent to a first oscillator and adjusting the current such that thefirst oscillator frequency approximates the frequency of an externalreference oscillator.
 32. A method for simultaneously calibrating aplurality of integrated circuit devices, comprising: supplying areference frequency to all devices; and commanding all devices toself-calibrate using the reference frequency.
 33. A system forfrequency-based calibration, comprising: a mechanism for linkingcalibration parameters to an internal frequency; and a mechanism forlinking the internal frequency to an external frequency.
 34. The systemof claim 33, wherein the mechanism for linking calibration parameters tothe internal frequency is a voltage controlled oscillator.
 35. Thesystem of claim 33, wherein the mechanism for linking the internalfrequency to an external frequency is a phase locked loop oscillator.36. A real time clock, comprising: a real time clock operating at afirst frequency; a wireless receiver for receiving a referencefrequency; and a mechanism for calibrating the real time clock frequencyusing the external reference frequency.
 37. An RFID system, comprising:an RFID tag implementing the system of claim 1; and an RFID reader incommunication with the RFID tag.
 38. An RFID system, comprising: an RFIDtag implementing the system of claim 20; and an RFID reader incommunication with the RFID tag.